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RF Power Amplifier Design
Markus Mayer & Holger Arthaber
Department of Electrical Measurements and Circuit Design Vienna University of Technology June 11, 2001
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RF Power Amplifier Design Markus Mayer & Holger Arthaber - - PowerPoint PPT Presentation
RF Power Amplifier Design Markus Mayer & Holger Arthaber - - PowerPoint PPT Presentation
RF Power Amplifier Design Markus Mayer & Holger Arthaber Department of Electrical Measurements and Circuit Design Vienna University of Technology June 11, 2001 Contents Basic Amplifier Concepts Class A, B, C, F, hHCA Linearity
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Efficiency Definitions
Drain Efficiency: Power Added Efficiency:
DC OUT D
P P = η
− ⋅ = − = G P P P
D DC IN OUT PA
1 1 η η
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Ideal FET Input and Output Characteristics
VGS IDS Im 2VP VP VDSmax VDD VK VDS V =V
GS P
V =0
GS
Ohmic Saturation Breakdown gm
DD K DD
V V V − = κ
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Maximum Output Power Match
VGS IDS Im 2VP VP VDSmax VDD VK VDS V =V
GS P
V =0
GS
Ohmic Saturation Breakdown gm m K DS OPT
I V V R − =
max
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Class A
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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Class A – Circuit
VDD RL
D G S
48% dB) 14 (e.g. 50%
PA
⋅ = = ⋅ = κ η κ η
A D
G G
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Class B
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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Class C
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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Class B and C – Circuit
VDD RL
D G S
f0
Class B Class C
% 65 dB) (8 6dB
- %
78
PA
⋅ = = ⋅ = κ η κ η
A D
G G % 1 % 100
PA →
→ → η η G
D
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Influence of Conduction Angle
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Class F (HCA ... harmonic controlled amplifier)
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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hHCA (half sinusoidally driven HCA)
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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Class F and hHCA – Circuit
VDD RL VDS ID
Ze(n)
0, n=even inf, n=even
Zo(n)
0, n=1 inf, n=odd
Class F hHCA
% 87 dB) (9 5dB
- 0%
10
PA
⋅ = = ⋅ = κ η κ η
A D
G G % 96 dB) (15 1dB 0% 10
PA
⋅ = + = ⋅ = κ η κ η
A D
G G
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hHCA – Third Harmonic Peaking
VGS IDS Im 2VP VP VDSmax VDD VK VDS VGS VDS 2p
p Q
IDS Im 2p
p Q
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Third Harmonic Peaking – Circuit
VDD RL
D G S
f0 3f0
% 87 dB) (14.6 0.6dB 91%
PA
⋅ = + = ⋅ = κ η κ η
A D
G G
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Linearity Aspects
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Linearity Aspects
Class AB Class C Class A Class B
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Linearity Aspects
Ideal strongly nonlinear model Strong-weak nonlinear model
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Amplifier Design – An Example
Balanced Amplifier Configuration
Port 1 Z=50 Ohm Port 2 Z=50 Ohm
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Amplifier Design – Simulation
Gate & Drain Waveforms
500 1000 1300 Time (ps)
Drain waveforms
- 5
5 10 15 20 25
- 1000
1000 2000 3000 4000 5000
Inner Drain Voltage (L, V) Amp Inner Drain Current (R, mA) Amp
500 1000 1300 Time (ps)
Gate waveforms
- 3
- 2
- 1
1
- 1000
- 500
500 1000
Inner Gate Voltage (L, V) Amp Inner Gate Current (R, mA) Amp
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Amplifier Design – Simulation
Dynamic Load Line & Power Sweep
3 6 9 12 15 Voltage (V)
Dynamic load line
- 2000
2000 4000 6000 8000
IVCurve (mA) IV_Curve Dynamic Load Line (mA) Amp
5 10 15 20 24 Power (dBm)
Power Sweep 1 Tone
10 20 30 40 10 20 30 40 50 60 70 80
Output Power (L, dBm) Amp PAE (R) Amp
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Amplifier Design – Measurements
Single Tone & Two Tone
5 1 1 5 2 2 5 3 3 5 4 5 1 1 5 2 2 5 3 3 5 P in [d B m ] P out [dBm], Gain [dB] 1 2 3 4 5 6 7 8 0 P A E [% ] P
- u
t G a in G a m m a In P A E 1 d B C P 10 20 30 40 50 60 5 10 15 20 25 30 35 P in [dB m ] P out [dBm], IMDD [dBc], Gain [dB] 10 20 30 40 50 60 P A E [% ] P
- ut
IM D D G ain P A E
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Amplifier Nonlinearity
Gain and Phase depends on Input Signal 3rd Order Gain-Nonlinearities:
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Amplifier Nonlinearity
Higher Output Level (close to Saturation) results in more Distortion/Nonlinearity
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Nonlinearity leads to?
Generation of Harmonics Intermodulation Distortion / Spectral Regrowth SNR (NPR) Degradation Constellation Deformation
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Intermodulation and Harmonics
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Spectral Regrowth
- 15
- 10
- 5
5 10 15
- 60
- 50
- 40
- 30
- 20
- 10
10 relative power / dB relative frequency / MHz ACPR1>60dB ACPR2>60dB ACPR1=16dB ACPR2=43dB
Energy in adjacent Channels ACPR (Adjacent Channel Leakage Power Ratio) increases
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Reduced NPR (Noise Power Ratio)
Output Signal of Nonlinear Amplifier Input Signal Degradation of Inband SNR „Noisy“ Constellation
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Constellation Deformation
Input Signal Output Signal of Nonlinear Amplifier (with Gain- and Phase-Distortion)
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Modeling of Nonlinearities
with Memory-Effects Volterra Series (=„Taylor Series with Memory“) without Memory-Effects Saleh Model Taylor Series Blum and Jeruchim Model AM/AM- and AM/PM-conversion
2 2 2
1 ) ( 1 ) ( r r r g r r r f
a a Θ Θ
+ = + = β α β α
better performance
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AM/AM- and AM/PM-Conversion
GaAs-PA
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AM/AM- and AM/PM-Conversion
LDMOS-PA
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How to preserve Linearity?
Backed-Off Operation of PA Simplest Way to achieve Linearity Linearity improving Concepts Predistortion Feedforward ...
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How to preserve Efficiency?
Efficiency improving Concepts Doherty Envelope Elimination and Restoration ... Linearity improving Concepts Higher Linearity at constant Efficiency Higher Efficiency at constant Linearity
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Direct (RF) Feedback
Classical Method Decrease of Gain Low Efficiency Feedback needs more Bandwidth than Signal Stability Problems at high Bandwidths
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Distortion Feedback
Feedback of outband Products only Higher Gain than RF feedback Stability Problems due to Reverse Loop
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Feedforward
Overcomes Stability Problem by forward-only Loops Critical to Gain/Phase-Imbalances 0.5dB Gain Error -31dB Cancellation 2.5° Phase Error -27dB Cancellation Well suited for narrowband application
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Cartesian Feedback
- 30
- 20
- 10
10 20 30
- 60
- 50
- 40
- 30
- 20
- 10
10 relative power / dB relative frequency / MHz
- riginal signal
predistorted signal
AM/AM- and AM/PM-correction High Feedback-Bandwidth Stability Problems
I Q I Q I Q modulator demodulator OPAs main amp. local
- scillator
RF-output baseband input
UMTS example:
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Digital Predistortion
Digital Implementation of „Cartesian Feedback“ Additional ADCs, DSP Power, Oversampling needed Loop can be opened no Stability Problems
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Analog Predistortion
Predistorter has inverse Function of Amplifier Leads to infinite Bandwidth (!) Hard to realize (accuracy)
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Analog Predistortion
Possible Realizations:
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LINC (Linear Amplification by Nonlinear Components)
AM/AM- and AM/PM-correction Digital separation required (accuracy!) High Bandwidth,
- versampling necessary
Stability guaranteed
signal separation s(t) s (t)
1
K K s (t)
2
K(s (t)+
1
s (t)) =Ks(t)
2
Ks (t)
1
Ks (t)
2
- 30
- 20
- 10
10 20 30
- 60
- 50
- 40
- 30
- 20
- 10
10 relative power / dB relative frequency / MHz ACPR1>60dB ACPR2>60dB ACPR1=18dB ACPR2=29dB s(t) s1(t)
UMTS example:
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Doherty Amplifier
Auxiliary amplifier supports main amplifier during saturation PAE can be kept high over a 6dB range
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Doherty Amplifier
Gain vs. Input Power No improvement of AM/AM- and AM/PM-distortion Behavior of auxiliary amplifier very hard (impossible) to realize Stability guaranteed Efficiency vs. Input Power
main amp. (A1)
- aux. amp. (A2)
PIN POUT doherty configuration (A1+A2)
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EER (Envelope Elimination and Restoration)
Separating phase and magnitude information Elimination of AM/AM-distortion Application of high-efficient amplifiers (independent of amplitude distortion) Stability guaranteed
signal separation amplitude information phase information RF input RF output high efficiency power amplifier
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EER (Envelope Elimination and Restoration)
Analog realization Limiter hard to build Accuracy problems Feedback necessary Digital realization Oversampling + high D/A- conversion rates required High power consumption
- f DSP and D/A-converters
Possible feedback elimination Compensation of AM/PM- distortion possible
peak detector supply voltage amplifier limiter high efficiency power amplifier RF output peak detector RF input D A D A D A amplitude information phase information modulator RF output high efficiency power amplifier digital signal processor local oscillator supply voltage amplifier I Q I Q digital baseband input
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EER (Envelope Elimination and Restoration)
Five times (!) oversampling necessary to achieve standard requirements Bandwidth of Magnitude- and phase-signal have higher than transmit signal
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- 20
- 10
10 20 30
- 60
- 50
- 40
- 30
- 20
- 10
10 relative power / dB relative frequency / MHz Magnitude Phase
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- 20
- 10
10 20 30
- 60
- 50
- 40
- 30
- 20
- 10
10 relative power / dB relative frequency / MHz ACPR1>60dB ACPR2>60dB ACPR1=33dB ACPR2=40dB ACPR1=51dB ACPR2=36dB ACPR1=53dB ACPR2=49dB full bandwidth 3⋅B0 bandwidth 5⋅B0 bandwidth 7⋅B0 bandwidth
UMTS example: UMTS example:
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Adaptive Bias
Varying/Switching of Bias-Voltage depending on Input Power Level Selection of Operating Point with high PAE Applicably for nearly each type of Amplifier
RF input peak detector bias control RF output high efficiency power amplifier
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Adaptive Bias
32 33 34 35 36 37 38 39 40 20 30 40 50 60 70 80 90
- utput power / dBm
power added efficiency / % VD=3.5V VD=4.5V VD=6.5V
Single tone PAE for switched VDD with VG kept constant Simply to implement Concept Stability guaranteed Possible problems: DC-DC converter with high efficiency necessary Possible Linearity Change (can increase and decrease) especially for HCAs
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Summary
Digital Realization required to achieve Accuracy Problem of Stability for high Bandwidth Application Higher Bandwidths (Oversampling) necessary, depending on Order of IMD cancellation Predistortion gives best Results while keeping Efficiency high (valid for high Output Levels > 40dBm)
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Figure References
- F. Zavosh et al,
“Digital Predistortion Techniques for RF Power Amplifiers with CDMA Applications”, Microwave Journal, Oct. 1999 Peter B. Kenington, “High-Linearity RF Amplifier Design”, Artech House, 2000 Steve C. Cripps, “RF Power Amplifiers for Wireless Communications”, Artech House, 1999
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