home page http://rubiola.org
Phase & Frequency Noise Metrology
- Introduction
- Measurement methods
- Microwave photonics
- Electronic and optical components
- AM noise and RIN
Outline
Enrico Rubiola
Phase & Frequency Noise Metrology Enrico Rubiola Outline - - PowerPoint PPT Presentation
Phase & Frequency Noise Metrology Enrico Rubiola Outline Introduction Measurement methods Microwave photonics Electronic and optical components AM noise and RIN home page http://rubiola.org Though frequency
Enrico Rubiola
2
Fourier frequency, Hz
102 103 104
(f )
2
dB[rad ]/Hz
Sα|ϕ
10 −164 −174 −204 −194 −184 Narda CNA 8596 s.no. 157 P instrument noise avg 10 spectra = 19 dBm single channel
ν0 = 9.2 GHz
Microwave circulator
3 Sϕ(f) = PSD of ϕ(t)
power spectral density
L(f) = 1 2Sϕ(f) dBc y(t) = ˙ ϕ(t) 2πν0 ⇒ Sy = f 2 ν2 Sϕ(f) σ2
y(τ) = E
1 2
2 .
Allan variance (two-sample wavelet-like variance) approaches a half-octave bandpass filter (for white noise), hence it converges for processes steeper than 1/f random fractional-frequency fluctuation random phase fluctuation
signal sources only
f h2f2 b0
2
ν0 f2/
x
2ln(2)h −1 )2 h−2 (2π 6 τ h0 /2τ
f−4 b−4 b−2f−2 b−1 f−1 h−2 f−2 h−1f−1 b−3f−3 Sϕ(f) Sy(f)
y 2
σ (τ)
white freq. white phase flicker phase.
f
white freq. flicker phase white phase
f
white phase flicker phase drift
τ
flicker freq. random walk freq. random flicker freq. random walk freq. white freq. flicker freq. walk freq.
h
freq.
h1
it is measured as Sϕ(f) = E {Φ(f)Φ∗(f)}
(expectation)
Sϕ(f) ≈ Φ(f)Φ∗(f)m
(average)
both signal sources and two-port devices
v(t) = Vp [1 + α(t)] cos [1 + ϕ(t)]
4
Any phase fluctuation can be converted into length fluctuation
L = 1 2π c ν0
Sϕ(f)
10–18 rad2/Hz @ 1 Hz
h−1 / f
SL(f)
L = 1 2π c ν0 h−1 / f
1.5x10–23 m2/Hz @ 1 Hz
f f τ
σ2 = 2 ln(2) h−1
4.5x10–12 m
σL(τ)
Any flicker spectrum h–1/f can be converted into a flat Allan variance
σ2
L = 2 ln(2) h−1
A residual flicker of –180 dBrad2/Hz at f = 1 Hz
b–1 = –180 dBrad2/Hz and ν0 = 10 GHz is equivalent to SL = 1.46x10–23 m2/Hz at f = 1 Hz
σ2 = 2x10–23 m2 thus σ = 4.5x10–12 m
for reference, the Bohr radius of the electron is R = 0.529 Å
6
DUT FFT
b(t) c(t) x = c–a y = c–b basics of correlation in practice, average on m realizations Syx(f) = E {Y (f)X∗(f)} = E {(C − A)(C − B)∗} = E {CC∗ − AC∗ − CB∗ + AB∗} = E {CC∗} Syx(f) = Scc(f) 0 as 1/√m Syx(f) = Y (f)X∗(f)m = CC∗ − AC∗ − CB∗ + AB∗m = CC∗m + O(1/m)
single-channel correlation
frequency S(f) 1/m
a(t), b(t) –> instrument noise c(t) –> DUT noise
Two separate instruments measure the same DUT. Only the DUT noise is common
phase noise measurements DUT noise, normal use a, b c instrument noise DUT noise background, ideal case a, b c = 0 instrument noise no DUT background, with AM noise a, b c ≠ 0 instrument noise AM-to-DC noise
7
! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=1 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=2 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=4 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=8 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=16 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=32 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=64 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=128 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=256 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=512 g=0.32
frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!
|Sxx| |Scc| |Re{Syx}| m=1024 g=0.32
frequency # #! #!! #!!! !%!# !%# # a v e r a g e d e v i a t i
|Re{Syx}| m
&'()*+,)-./0!+)1!##!#!$2!!3#4!05+6)789 :%6;5'<(0=*0,/*$!!>
|Re{Syx}| with C≠0,
Increasing m: first, Syx decreases => single-channel noise rejection then, Sxx shrinks => increased confidence level
8
Application to AM/PM noise: E. Rubiola, V. Giordano, Rev. Sci. Instrum. 71(8) p.3085-3091, Aug 2000
0º 0º 0º 180º T2 A B X = A + B X = A – B T1
Syx = k (T2 – T1) / 2
X and Y are uncorrelated The cross spectrum is proportional to the temperature difference
9
Invented by J. Hall for gas spectroscopy. The gain is increased by the number of times the light beam circulates in the cavity
gas cell DUT
Also works with RF/microwave carrier, provided the DUT be “transparent”. For small no. of roundtrips, gives the appearance of “real-time”
10
fluctuating error δZ => noise sidebands ℜ{δZ} => AM noise x(t) cos(ω0t) ℑ{δZ} => PM noise –y(t) sin(ω0t)
amplification before detecting
Derives from H. Sann, MTT 16(9) 1968, and F . Labaar, Microwaves 21(3) 1982 Later, E. Ivanov, MTT 46(10) oct 1998, and Rubiola, RSI 70(1) jan 1999
0º –90º FFT
x(t) y(t)
pump
bridge
(microwave) error amplifier
(t)
DUT
Z
V0 cos(0t)
Z
x(t) cos(0t) – y(t) sin(0t)
–
hybrid junction
11
fixes the arbitrary phase, gain asymmetry and quadrature defect
eliminates the amplifier noise. Works with a single amplifier!
Re Im Up Dn
v(t)/2 v(t)/2 v(t) null fluct
Sud(f) = 1 2
integr matrix D R0=50 Ω matrix B matrix G v2 w1 w2 matrix B matrix G w1 w2 FFT analyz. atten atten
x t ( )
Q I I−Q modul
’
γ’ atten Q I I−Q detect RF LO Q I I−Q detect RF LO g ~ 40dB g ~ 40dB v1 v2 v1 u1 u2 z2 z1 atten DUT γ Δ’
R R
10−20dB coupl. power splitter pump channel a channel b (optional) rf virtual gnd null Re & Im RF suppression control manual carr. suppr. pump LO diagonaliz. readout readout arbitrary phase
automatic carrier arbitrary phase pump
I−Q detector/modulator G: Gram Schmidt ortho normalization B: frame rotation
inner interferometer
CP1 CP2 CP3 CP4
−90° 0° I Q RF LO
12
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+ f
,-./0
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S 1 + f
,-(2,33./0
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S"
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89:;*)32(6 P
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S , & f
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#
S"
1//////////'(2)*+3
N
456-78-/0-8968:;<=/*+
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1!!$MEC3 1!!%MEC3 1!!>MEC3 1!!CMEC3 1!!DMEC3
Noise of a pair of HH-109 hybrid junctions Background noise of the fixed-value bridge (larger m) Background noise of the fixed-value bridge
!"# !"$ %&''() *+,-)(./'0
1&2'+('.3'(42(,%56.78
9 : f
;<'/;..=78
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S! : f
;<=78
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N
P
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>!!KKCM? >!!JKCM? >!!IKCM? ! >!!FKCM?
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>!!MKCM?
plot 459
Noise of a by-step attenuator
Averaged spectra must be smooth Average on m spectra: confidence of a point improves by 1/m1/2 interchange ensemble with frequency: smoothness 1/m1/2
13
14
(dc circuits not shown)
15
1 10
3 2 4 5
−180 10 10 10 −140 −170 −160
i n t e r f e r
e t e r
Fourier frequency, Hz
−220 −210
s a t u r a t e d m i x e r c
r e l . s a t . m i x . double interf. interferometer residual flicker, by−step interferometer residual flicker, fixed interferometer residual flicker, fixed interferometer residual flicker, fixed interferometer, ±45° detection
S (f)
dBrad2/Hz ϕ
real−time
nested interferometer mixer, interferometer
saturated mixer d
b l e i n t e r f e r
e t e r
−200 −190 10 −150 measured floor, m=32k
expected, m=4x106
16
mixer saturated at both inputs
conceptually incorrect
microwave gain
17
x t) ω ysin( − V cos( t) ω modulated signal carrier modulation V cos( t) ω V0
=
x
α
cos( t) ω x
B
carrier modulation in−phase amplitude−modulated signal V0
ϕ=
y V cos( t) ω t) ω ysin( −
C
carrier
ϕ
quadrature modulation phase−modulated signal cos(
A
t) ω
I−Q modulator V cos( t) ω cos( t) ω x t) ω ysin( − V cos( t) ω + LO I Q RF I−Q modul
y x
cos( t) ω x t) ω ysin( − RF output RF input carrier modulation sidebands 90°
RF I Q LO pump
P0 = power of the carrier Px = power of the in-phase sidebands Py = power of the quadrature sidebands
αrms =
ϕrms =
AM and PM can be defined as
18
−90° − 9 ° −90° − 9 °
0° 0° 0° 1 8 ° −90° −90° 0° ° 0° 0° 180° °
Q I fine carrier control
0° 0° 0° 1 8 °
Q I I−Q detect
g
v2a matrix R v1a w1a w2a
LO RF
ampli & det. readout channel a Q I I−Q detect
g
v2b matrix R
1b
v w1b w2b
LO RF
ampli & detector readout channel b virtual gnd RF Δ" dual channel FFT analyzer ( ) DUT Δ’ γ inner interferometer
−20 dB by−pass CP1 CP2 CP3 CP4
Q I I−Q modul
1
u
2
u t dt z matrix D
2
z
1
z
LO RF
automatic carrier control matrix lock−in in
amplifier to AM/PM
Σ Σ
AM PM modulation input from AM out AM out power modul input meter
Light blue: work in progress The dual-bridge contains almost all the blocks needed to calibrate the measurement
20 10 GHz, 10 μs
fiber: attenuation 0.2 dB/km, thermal coeff. 6.8 10-6/K cable: attenuation 0.8 dB/m, thermal coeff. ~ 10-3/K
Rubiola, Salik, Huang, Yu, Maleki, JOSA-B 22(5) p.987–997 (2005)
Φ(s) = Hϕ(s)Φi(s)
Laplace transforms
Sy(f) = |Hy(f)|2 Sϕ i(s)
|Hϕ(f)|2 = 4 sin2(πfτ) |Hy(f)|2 = 4ν2 f 2 sin2(πfτ) 10 GHz, 10 μs
−
Σ
kϕ −s
e
τ Φo(s) Φi(s) V
= Φo(s)
τ −s
(1−e )Φi(s) = mixer
+
detector
mW 10
Pλ τd = 1.. 100 µ s
EOM
90° adjust τ∼ _0 laser µm 1.55
mW 100
_ ∼ τd 0 20−40 dB R0 52 dB FFT analyz. (t) vo
(0.2−20 km) power ampli input microwave (calib.) phase
The short arm can be a microwave cable or a photonic channel Laplace transforms
Qeq = πν0τ
21
blue and black plots overlap magenta, red, green => instrument noise blue, black => noise of the sapphire
(frequency flicker) of a 10 GHz sapphire
necessary
Att FFT
DC JDS Uniphase JDS Uniphase 1,5 µm = Contrôleur de polarisation Photodiode DSC40S Déphaseur Ampli DC Analyseur FFT (HP 3561A)
Coupleur 10 dB
Ampli RF
3dB
Ampli AML 8-12GHz
LO RF 5 dBm 10 dBm
ISO ISO
Fibre 2 Km
laser EOM SiGe ampli phase 2 km sapphire oscillator
Volyanskiy & al., JOSAB (in press). Also arXiv:0807.3494v1 [physics.optics] July 2008.
22
Improvements
Derives from: E. Salik, N. Yu, L. Maleki, E. Rubiola, Proc. Ultrasonics-FCS Joint Conf., Montreal, Aug 2004 p.303-306
Volyanskiy & al., JOSAB (in press) and arXiv:0807.3494v1 [physics.optics] July 2008.
!'" !(" !%" !#" !"" '" (" %" #" ;<9/0=.*17.1>*-?@17.1A=9B.19C.011-/1*.@9*717.1!"DB12E?>*.1#FG5 6A.01H.*IE<.J1K!"F3419C.01-/1*.@9*717.1#"DB12E?>*.1%FG51 J.Cussey 20/02/07 Mesure200avg.txt
–20 –180 –40 –60 –80 –160 –140 –120 –100 101 102 103 104 105
Fourier frequency, Hz S(f), dBrad2/Hz residual phase noise (cross-spectrum), short delay (0), m=200 averaged spectra, unapplying the delay eq. with =10 s (2 km)
J.Cussey, feb 2007
y = 10–12 baseline
FFT average effect FFT average effect FFT average effect
23
Σ
+ + A
free noise
Vo(s) V
i(s)
initial conditions, noise, or locking signal βf(s) selector βd
τd
e−s = (s) delay
τd
e−s model output
true in practice, delay + selector delay = (s) β
+2π/τd . . . . H(s) . . . . σ l=+3 l=+2 l=+1 l=0 l=−1 l=−2 l=−3 . . . . . . . . j ω τd ln(A) 1 +6π/τd −6π/τd +4π/τd −2π/τd −4π/τd
1 2 3 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20
f * tau transfer function |H(jf)|^2 A=1 A=0.75 A=0.71 A=0.65 A=0.50 A=0.30
file le−calc−hdly−flt src allplots−leeson
delay−line loop, no selection filter
General feedback theory H (s) = Vo(s) Vi(s) = 1 1 A (s) Delay-line oscillator H (s) = 1 1 A e
sτd
Location of the roots s
l = 1
ln(A ) + j2
l integer l( , )
Barkhausen condition for oscillation: Aß = 1
24
1000 10000 1e+05 1e+06 0.01 0.1 1 10 100 1000 10000 1e+05 1e+06 1e+07 1e+08
phase noise |H(jf)|^2 delay−line oscillator with selector
frequency, Hz
parameters: tau = 2E−5 s m = 2E5 nu_m = 10 GHz Q = 2000
file le−calc−dly−hphase src allplots−leeson
1
Σ
+ +
(s) Ψ (s) Φ free noise (s) B
f
τ 1 / (1+s ) = (s) B
τd
e−s = delay (s) B
τd
e−s = delay phase noise input phase noise
selector in practice, delay + selector ω m σ µ=0 µ=−1 µ=−2 . . . . . . . . µ=+2 µ=+1 = j ωm 2Q µ m τd Δω = − 2Q µ m τd Δω = − 2Q ωm (2π/τd) 2Q2 µ m
2
τd σ = − (2π/τd) µ (2π/τd) µ
General feedback theory H(s) = Φ(s) Ψ(s) = 1 1 − B(s) Delay-line oscillator H(s) = 1 + sτf 1 + sτf − e−sτd Location of the roots sµ = −2Q2 τd µ m 2 + j 2π τd µ − 2Q τd µ m
25 fL = 1 4π2τ 2 fL = ν0 2Q Qeq = πν0τ Qeq=3×105 ← L=4km Sϕ(f) ≃ f 2
L
f 2 Sψ(f) for f ≪ fL fL=8kHz
Leeson formula
10–11 Allan deviation h−1 = b−3/ν2 6.3×10–24 8.8×10–24 σ2
y = 2 ln(2) h−1
b–3 = 6.3×10–4 (–32 dB)
26 expected phase noise b–3 ≈ 6.3×10–4 (–32 dB)
b–3=10–3 (–30dB)
Agilent E8257c, 10 GHz, low-noise opt. Wenzel 501-04623 OCXO 100 MHz
101 102 103 104 105 –20 –40 –60 –80 –160 –140 –120 –100
S(f), dBrad2/Hz
Phase noise of the opto-electronic oscillator (4 km)
frequency, Hz
OEO: Kirill Volyanskiy, may 2007
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D==
Kiryll Volyanskiy, jan 2008
27
the mode selector filter
(AML SiGe parallel amplifiers exhibits lowest flicker, but low have gain 22 dB)
The oscillator phase noise minima are 6 dB lower than b0=N/P0 (white noise) m = 0.725 (Prf=11 dBm) (Sφ)min = –142 dB F = 10 dB (incl. couplers) η = 0.6 νl = 194 THz (Sϕ)min = 8 m2
R0 hνl qη 2 1 P
2 l
+ 2 hνl η 1 P l
1
10
2
10
3
10
4
10
5
160 140 120 100 80 60 40 20 Frequency (Hz) S (dB rad2/Hz) 8dBm 9dBm 11dBm
b–2 = –50.5 dBrad2/Hz (8 dBm) b–3 = –26 dBrad2/Hz b–4 = –7 dBrad2/Hz b–2 = –53 dBrad2/Hz (9 dBm) b–2 = –57 dBrad2/Hz (11 dBm) file 923-kirill-oeo E.R & K.Voliansky, jan 2008 (S)min = –142 dBrad2/Hz (11 dBm) the power is measured at the amplifier output
we get P0 = 6.4 μW (-22 dBm) Pl ≈ 0.71 mW There is room for engineering
Prf is given, thus V0 =(2RP)1/2 Vπ is estimated (4.5 V at 10 GHz) Use m = 2J1 πV0 Vπ
Vp, V πV0/Vπ m 11 1.122 0.860 0.783 9 0.891 0.683 0.644 8 0.794 0.6.09 0.581
Feeding the available data in the model Get 28
30
stopband
stopband
near-dc flicker
no carrier
S(f) f S(f) f
noise up-conversion
near- dc noise
expand and select the ω0 terms carrier
vi(t) = Vi ejω0t + n′(t) + jn′′(t)
non-linear (parametric) amplifier
vo(t) = Vi
get AM and PM noise
α(t) = 2 a2 a1 n′(t) ϕ(t) = 2 a2 a1 n′′(t)
The AM and the PM noise are independent of Vi , thus of power
vo(t) = a1vi(t) + a2v2
i (t) + . . . substitute
(careful, this hides the down-conversion)
the parametric nature of 1/f noise is hidden in n’ and n”
ω0 = ?
no flicker
ω0
The noise sidebands are proportional to the input carrier
near-dc noise
There is also a linear parametric model, which yields the same results
31
=−10dBm Pin=−5dBm Pin=0dBm −165 −160 −155 −150 −145 −140 −135 −130 −125 −120 f, Hz Phase noise, dBrad 2/Hz SiGe LPNT32
bias 2V, 20 mA
=−15dBm
100 1 10 1000 10000 100000 −175 P −170
in
Pin
2
23 Jan 07 Avantek UTC573, 10 MHz 56 dB
NMS floor
f, Hz
120 130 140 150 160 170 180 1 10 100 1000 10000 100000
1Amp Pin=5dBm 3Amps Pin= 24dBm 2Amps Pin= 5dBm
power
inverse of the carrier power
because fc depends on the input power
3 dB, with 2 amplifiers 5 dB, with 3 amplifiers
S (f), dB.rad /Hz
2
Pin= −5 dBm Single amplifier Pin= −6 dBm JS2, 10 GHz 15 Feb 2007
f, Hz
50 70 90 110 130 150 170 1 10 100 1000 10000 100000
Phase noise vs. power Phase noise of cascaded amplifiers Phase noise of paralleled amplifiers
noise flicker is expected to decrease by 3 dB
Regenerative amplifiers
the noise source at each roundtrip is correlated
32
r(t) iso iso P
90° ° 0° 90°
Pµ
RF LO IF
FFT analyz. power meter
=6dB (detection of or )
PLL
synth. 9.9GHz MHz 100 power ampli laser YAG 1.32 µ m
EOM
(3dBm)
s(t)
hybrid
g=37dB g’=52dB (carrier suppression)
phase & aten.
50% coupler
infrared v(t) 22dBm
(13dBm)
monitor
photodiodes under test microwave neardc
The noise of the ∑ amplifier is not detected [Rubiola, Salik, Yu, Maleki, Electron. Lett. 39(19) p.1389-1390 (2003) ]
photodiode Sα(1 Hz) Sϕ(1 Hz) estimate uncertainty estimate uncertainty HSD30 −122.7
−7.1 +3.4
−127.6
−8.6 +3.6
DSC30-1K −119.8
−3.1 +2.4
−120.8
−1.8 +1.7
QDMH3 −114.3
−1.5 +1.4
−120.2
−1.7 +1.6
unit dB/Hz dB dBrad2/Hz dB Rubiola, Salik, Yu, Maleki, MTT 54(2) p.816-820, Feb 2006
33
AM and PM 1/f noise
photodetector 1/f noise
in order to take the full benefit from the low noise of the junction
W: waving a hand 0.2 m/s, 3 m far from the system B: background noise P: photodiode noise S: single spectrum, with optical connectors and no isolators B: background noise P: photodiode noise A: average spectrum, with optical connectors and no isolators B: background noise P: photodiode noise F: after bending a fiber, 1/f noise can increase unpredictably B: background noise P: photodiode noise
Rubiola, Salik, Yu, Maleki, MTT 54(2) p.816-820, Feb 2006
34
f
Sϕ(f)
10–32 @ 1 Hz
τ
4 x 1 0–17 /
f
10–12 rad2/Hz @ 1 Hz
Sy(f)
Sy(f) = f 2 ν2 Sϕ(f) σ2
y(τ) = [1.038 + 3 ln(2πfHτ)]
h1 (2π)2 1 τ 2 b−1/f h1f
limited by the 1/f phase noise of a single component, –120 dBrad2/Hz @ f=1 Hz
yields σy(τ) = 4x10–16/τ
36
arXiv:physics/0512082v1 [physics.ins-det]
monitor source under test dual channel FFT analyzer vb va Pb Pa power meter monitor R0 R0 Pa Pb
coupler
power meter
coupler
source under test R R va vb dual channel FFT analyzer power meter microwave
monitor dc
dc power meter
coupler coupler
source under test Pb Pa R R dual channel FFT analyzer va vb monitor
−123.1 10 102 103 104 105
Fourier frequency, Hz
avg 2100 spectra = −10.2 dBm P
Wenzel 501−04623E 100 MHz OCXO
(f ) Sα
dB/Hz −163.1 −153.1 −143.1 −133.1
feeding the same signal into the phase detector
enables to validate the instrument
AM noise of RF/microwave sources Laser RIN AM noise of photonic RF/microwave sources
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20mA 30mA 40mA 60mA 80mA 100mA
Kirill Volyanskiy
37 source h−1 (flicker) (σα)floor Anritsu MG3690A synthesizer (10 GHz) 2.5×10−11 −106.0 dB 5.9×10−6 Marconi synthesizer (5 GHz) 1.1×10−12 −119.6 dB 1.2×10−6 Macom PLX 32-18 0.1 → 9.9 GHz multipl. 1.0×10−12 −120.0 dB 1.2×10−6 Omega DRV9R192-105F 9.2 GHz DRO 8.1×10−11 −100.9 dB 1.1×10−5 Narda DBP-0812N733 amplifier (9.9 GHz) 2.9×10−11 −105.4 dB 6.3×10−6 HP 8662A no. 1 synthesizer (100 MHz) 6.8×10−13 −121.7 dB 9.7×10−7 HP 8662A no. 2 synthesizer (100 MHz) 1.3×10−12 −118.8 dB 1.4×10−6 Fluke 6160B synthesizer 1.5×10−12 −118.3 dB 1.5×10−6 Racal Dana 9087B synthesizer (100 MHz) 8.4×10−12 −110.8 dB 3.4×10−6 Wenzel 500-02789D 100 MHz OCXO 4.7×10−12 −113.3 dB 2.6×10−6 Wenzel 501-04623E no. 1 100 MHz OCXO 2.0×10−13 −127.1 dB 5.2×10−7 Wenzel 501-04623E no. 2 100 MHz OCXO 1.5×10−13 −128.2 dB 4.6×10−7 worst best