Phase & Frequency Noise Metrology Enrico Rubiola Outline - - PowerPoint PPT Presentation

phase frequency noise metrology
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Phase & Frequency Noise Metrology Enrico Rubiola Outline - - PowerPoint PPT Presentation

Phase & Frequency Noise Metrology Enrico Rubiola Outline Introduction Measurement methods Microwave photonics Electronic and optical components AM noise and RIN home page http://rubiola.org Though frequency


slide-1
SLIDE 1

home page http://rubiola.org

Phase & Frequency Noise Metrology

  • Introduction
  • Measurement methods
  • Microwave photonics
  • Electronic and optical components
  • AM noise and RIN

Outline

Enrico Rubiola

slide-2
SLIDE 2

2

Fourier frequency, Hz

102 103 104

(f )

2

dB[rad ]/Hz

Sα|ϕ

10 −164 −174 −204 −194 −184 Narda CNA 8596 s.no. 157 P instrument noise avg 10 spectra = 19 dBm single channel

ν0 = 9.2 GHz

Lower phase noise is required Though frequency standards are moving to optics (and beyond), RF and microwaves are inevitable

Microwave circulator

slide-3
SLIDE 3

Phase noise & friends

3 Sϕ(f) = PSD of ϕ(t)

power spectral density

L(f) = 1 2Sϕ(f) dBc y(t) = ˙ ϕ(t) 2πν0 ⇒ Sy = f 2 ν2 Sϕ(f) σ2

y(τ) = E

1 2

  • yk+1 − yk

2 .

Allan variance (two-sample wavelet-like variance) approaches a half-octave bandpass filter (for white noise), hence it converges for processes steeper than 1/f random fractional-frequency fluctuation random phase fluctuation

signal sources only

f h2f2 b0

2

ν0 f2/

x

2ln(2)h −1 )2 h−2 (2π 6 τ h0 /2τ

f−4 b−4 b−2f−2 b−1 f−1 h−2 f−2 h−1f−1 b−3f−3 Sϕ(f) Sy(f)

y 2

σ (τ)

white freq. white phase flicker phase.

f

white freq. flicker phase white phase

f

white phase flicker phase drift

τ

flicker freq. random walk freq. random flicker freq. random walk freq. white freq. flicker freq. walk freq.

h

freq.

h1

it is measured as Sϕ(f) = E {Φ(f)Φ∗(f)}

(expectation)

Sϕ(f) ≈ Φ(f)Φ∗(f)m

(average)

both signal sources and two-port devices

v(t) = Vp [1 + α(t)] cos [1 + ϕ(t)]

slide-4
SLIDE 4

Mechanical stability

4

Any phase fluctuation can be converted into length fluctuation

L = 1 2π c ν0

Sϕ(f)

10–18 rad2/Hz @ 1 Hz

h−1 / f

SL(f)

L = 1 2π c ν0 h−1 / f

1.5x10–23 m2/Hz @ 1 Hz

f f τ

σ2 = 2 ln(2) h−1

4.5x10–12 m

σL(τ)

Any flicker spectrum h–1/f can be converted into a flat Allan variance

σ2

L = 2 ln(2) h−1

A residual flicker of –180 dBrad2/Hz at f = 1 Hz

  • ff the 10 GHz carrier is equivalent to

b–1 = –180 dBrad2/Hz and ν0 = 10 GHz is equivalent to SL = 1.46x10–23 m2/Hz at f = 1 Hz

σ2 = 2x10–23 m2 thus σ = 4.5x10–12 m

for reference, the Bohr radius of the electron is R = 0.529 Å

  • Don’t think “this is just engineering” !!!
  • Learn from non-optical microscopy (bulk matter, 5x10–14 m)
  • Careful DC section (capacitance and piezoelectricity)
  • The best advice is to be at least paranoiac
slide-5
SLIDE 5

1 – Measurement methods

slide-6
SLIDE 6

Correlation measurements

6

DUT FFT

b(t) c(t) x = c–a y = c–b basics of correlation in practice, average on m realizations Syx(f) = E {Y (f)X∗(f)} = E {(C − A)(C − B)∗} = E {CC∗ − AC∗ − CB∗ + AB∗} = E {CC∗} Syx(f) = Scc(f) 0 as 1/√m Syx(f) = Y (f)X∗(f)m = CC∗ − AC∗ − CB∗ + AB∗m = CC∗m + O(1/m)

single-channel correlation

frequency S(f) 1/m

a(t), b(t) –> instrument noise c(t) –> DUT noise

Two separate instruments measure the same DUT. Only the DUT noise is common

phase noise measurements DUT noise, normal use a, b c instrument noise DUT noise background, ideal case a, b c = 0 instrument noise no DUT background, with AM noise a, b c ≠ 0 instrument noise AM-to-DC noise

  • E. Rubiola, The magic of cross-spectrum measurements from DC to optics, http://rubiola.org
slide-7
SLIDE 7

Cross-spectrum, increasing m

7

! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=1 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=2 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=4 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=8 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=16 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=32 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=64 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=128 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=256 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=512 g=0.32

frequency ! "! #!! #"! $!! !%!!# !%!# !%# # #!

|Sxx| |Scc| |Re{Syx}| m=1024 g=0.32

frequency # #! #!! #!!! !%!# !%# # a v e r a g e d e v i a t i

  • n

|Re{Syx}| m

&'()*+,)-./0!+)1!##!#!$2!!3#4!05+6)789 :%6;5'<(0=*0,/*$!!>

  • E. Rubiola, The magic of cross-spectrum measurements from DC to optics, http://rubiola.org

|Re{Syx}| with C≠0,

Increasing m: first, Syx decreases => single-channel noise rejection then, Sxx shrinks => increased confidence level

slide-8
SLIDE 8

The thermal noise is rejected as any signal. The limit Sφ = P0/kT does not apply

8

  • C. M. Allred, A precision noise spectral density comparator, J. Res. NBS 66C no.4 p.323-330, Oct-Dec 1962

Application to AM/PM noise: E. Rubiola, V. Giordano, Rev. Sci. Instrum. 71(8) p.3085-3091, Aug 2000

0º 0º 0º 180º T2 A B X = A + B X = A – B T1

Syx = k (T2 – T1) / 2

X and Y are uncorrelated The cross spectrum is proportional to the temperature difference

slide-9
SLIDE 9

Carrier recirculation

9

Invented by J. Hall for gas spectroscopy. The gain is increased by the number of times the light beam circulates in the cavity

gas cell DUT

Also works with RF/microwave carrier, provided the DUT be “transparent”. For small no. of roundtrips, gives the appearance of “real-time”

slide-10
SLIDE 10

Bridge (interferometric) method

10

fluctuating error δZ => noise sidebands ℜ{δZ} => AM noise x(t) cos(ω0t) ℑ{δZ} => PM noise –y(t) sin(ω0t)

  • Carrier suppression => the error amplifier cannot flicker: it does know ω0
  • High gain, due to the (microwave) error amplifier
  • Low noise floor => the noise figure of the (microwave) error amplifier
  • High immunity to the low-frequency magnetic fields due to the microwave

amplification before detecting

  • Rejection of the master oscillator’s noise
  • Detection is a scalar product => signal-processing techniques

Basic ideas

Derives from H. Sann, MTT 16(9) 1968, and F . Labaar, Microwaves 21(3) 1982 Later, E. Ivanov, MTT 46(10) oct 1998, and Rubiola, RSI 70(1) jan 1999

0º –90º FFT

x(t) y(t)

pump

bridge

(microwave) error amplifier

(t)

DUT

Z

V0 cos(0t)

Z

x(t) cos(0t) – y(t) sin(0t)

  • +

hybrid junction

slide-11
SLIDE 11

Actual block diagram

  • E. Rubiola, V. Giordano, Rev. Sci. Instrum. 73(6) pp.2445-2457, June 2002

11

  • Coarse and fine carrier suppression reduces the flicker noise
  • Scalar product gives v1(t) and v2(t) in Cartesian frame. Linear algebra

fixes the arbitrary phase, gain asymmetry and quadrature defect

  • Closed-loop control of the carrier suppression works as a RF VGND
  • Correlation is possible, using two amplifiers and two detectors
  • Correlating the signals detected on two orthogonal axes (±45º)

eliminates the amplifier noise. Works with a single amplifier!

Concepts

Re Im Up Dn

v(t)/2 v(t)/2 v(t) null fluct

Sud(f) = 1 2

  • Sα(f) − Sϕ(f)
  • dual

integr matrix D R0=50 Ω matrix B matrix G v2 w1 w2 matrix B matrix G w1 w2 FFT analyz. atten atten

x t ( )

Q I I−Q modul

γ’ atten Q I I−Q detect RF LO Q I I−Q detect RF LO g ~ 40dB g ~ 40dB v1 v2 v1 u1 u2 z2 z1 atten DUT γ Δ’

R R

10−20dB coupl. power splitter pump channel a channel b (optional) rf virtual gnd null Re & Im RF suppression control manual carr. suppr. pump LO diagonaliz. readout readout arbitrary phase

  • var. att. & phase

automatic carrier arbitrary phase pump

I−Q detector/modulator G: Gram Schmidt ortho normalization B: frame rotation

inner interferometer

CP1 CP2 CP3 CP4

−90° 0° I Q RF LO

slide-12
SLIDE 12

Example of results

12

!"# !"$ !"% !"& '())*+

!, -

f

./)0.11234

$

S

  • f

./234

",

S

5678+*10)9

N

:1111111111111./9234; )< P

"=1!%>!1./9

0?81&$@15A*'B)0 BC(1)<1'D077> 078+*1E7'0+> 0)91!0F1!G k T

B

2P =1!!HH>!1./:)0.11;234

$ " "

:!!#I>%; :!!II>%; :!!JI>%; :!!HI>%; :!!KI>%;

!!J"># !!H"># !!K"># !$""># !$!">#

L(E)6*)1<)*ME*7'NF134

!"# !"$ !"% &'(()*

+ f

,-./0

!1

S 1 + f

,-(2,33./0

#

S"

(4 53333333333,-6./07

N

89:;*)32(6 P

"

2<;3=#38>)&?(2 @3!%A!3,-6 ?B'3(43&C2::A 2:;*)3D:&2*A 2(63!2E3!F

G'D(9)(34()HD):&IE3/0

k T

B

.P @3!!JJA!3,-5(2,337./0

# " "

!"K !!L"AK !!M"AK !!J"AK !!="AK !#""AK

5!!LLA%7 5!!MLA%7 5!!JLA%7 !

!"

5!!%LA%7 5!!KLA%7

!"# !"$ !"%

& f

'()*+

!,

S , & f

'(-.'//)*+

#

S"

1//////////'(2)*+3

N

456-78-/0-8968:;<=/*+

!"> P

"

?7:@A8/.-2 7:?B-628:B/:57?8 !!CDE$ !!DDE$ ! !" .F@/!G/?H8;B-. BI5/-0/;J.::E .:@A8/6:;.AE .-2/!.=/!K L/!DED/'(2 !!>DE$ !!GDE$ !!MDE$

1!!$MEC3 1!!%MEC3 1!!>MEC3 1!!CMEC3 1!!DMEC3

Noise of a pair of HH-109 hybrid junctions Background noise of the fixed-value bridge (larger m) Background noise of the fixed-value bridge

!"# !"$ %&''() *+,-)(./'0

1&2'+('.3'(42(,%56.78

9 : f

;<'/;..=78

$

S! : f

;<=78

"9

S

'3 >..........;<0=78?

N

P

"

@A&.'3.%B/,,C /,-)(.2,%/)C /'0.!/6.!D /E-.#F!.*G(%@'/ H.!#CI.;<0 k T

" B

=P

" H.!!IJCI.;<>'/;..?=78 $

!!K"C$ !!J"C$ !!I"C$ !!L"C$ !$""C$

>!!KKCM? >!!JKCM? >!!IKCM? ! >!!FKCM?

!"

>!!MKCM?

plot 459

Noise of a by-step attenuator

Averaged spectra must be smooth Average on m spectra: confidence of a point improves by 1/m1/2 interchange ensemble with frequency: smoothness 1/m1/2

slide-13
SLIDE 13

The complete machine (100 MHz)

13

slide-14
SLIDE 14

A 10 GHz experiment

14

(dc circuits not shown)

slide-15
SLIDE 15

15

1 10

3 2 4 5

−180 10 10 10 −140 −170 −160

i n t e r f e r

  • m

e t e r

  • correl. saturated mixer

Fourier frequency, Hz

−220 −210

s a t u r a t e d m i x e r c

  • r

r e l . s a t . m i x . double interf. interferometer residual flicker, by−step interferometer residual flicker, fixed interferometer residual flicker, fixed interferometer residual flicker, fixed interferometer, ±45° detection

S (f)

dBrad2/Hz ϕ

real−time

  • correl. & avg.

nested interferometer mixer, interferometer

saturated mixer d

  • u

b l e i n t e r f e r

  • m

e t e r

−200 −190 10 −150 measured floor, m=32k

Comparison of the background noise

expected, m=4x106

slide-16
SLIDE 16

16

AM-PM calibration

  • In most cases the phase detector is a double-balanced

mixer saturated at both inputs

  • In a double-balanced mixer, the offset is affected by power,
  • AM noise is detected as it was PM noise, which is

conceptually incorrect

  • kφ/kα can be as low as 5
  • In the bridge, the effect of AM noise is divided by the

microwave gain

slide-17
SLIDE 17

17

Primary AM-PM calibration

x t) ω ysin( − V cos( t) ω modulated signal carrier modulation V cos( t) ω V0

=

x

α

cos( t) ω x

B

carrier modulation in−phase amplitude−modulated signal V0

ϕ=

y V cos( t) ω t) ω ysin( −

C

carrier

ϕ

quadrature modulation phase−modulated signal cos(

A

t) ω

I−Q modulator V cos( t) ω cos( t) ω x t) ω ysin( − V cos( t) ω + LO I Q RF I−Q modul

y x

  • modul. input

cos( t) ω x t) ω ysin( − RF output RF input carrier modulation sidebands 90°

  • utput

RF I Q LO pump

P0 = power of the carrier Px = power of the in-phase sidebands Py = power of the quadrature sidebands

αrms =

  • Px/P0

ϕrms =

  • Py/P0
  • Generate reference AM and PM by adding sidebands to a carrier
  • Power detectors provide absolute reference of PM as a null of AM
  • Accurate (0.02 dB) power-ratio measurement with commercial power-meters
  • Correct IQ modulators and detectors with linear algebra (2x2 matrices)
  • Transfer the accuracy of a LF (kHz range) lock-in amplifier to RF/microwave
  • Worst-case accuracy 0.3 dB => improvement in progress

Basic ideas

AM and PM can be defined as

  • E. Rubiola – unpublished
slide-18
SLIDE 18

18

Bridge (interferometric) instrument

−90° − 9 ° −90° − 9 °

0° 0° 0° 1 8 ° −90° −90° 0° ° 0° 0° 180° °

Q I fine carrier control

0° 0° 0° 1 8 °

Q I I−Q detect

g

v2a matrix R v1a w1a w2a

LO RF

ampli & det. readout channel a Q I I−Q detect

g

v2b matrix R

1b

v w1b w2b

LO RF

ampli & detector readout channel b virtual gnd RF Δ" dual channel FFT analyzer ( ) DUT Δ’ γ inner interferometer

−20 dB by−pass CP1 CP2 CP3 CP4

Q I I−Q modul

1

u

2

u t dt z matrix D

2

z

1

z

LO RF

automatic carrier control matrix lock−in in

  • sc

amplifier to AM/PM

Σ Σ

AM PM modulation input from AM out AM out power modul input meter

  • E. Rubiola, V. Giordano, Rev. Sci. Instrum 73 6 p.2445–57, jun 2002. Also arXiv:physics/0503015

Light blue: work in progress The dual-bridge contains almost all the blocks needed to calibrate the measurement

slide-19
SLIDE 19

2 – Microwave photonics

slide-20
SLIDE 20

Opto-electronic discriminator

20 10 GHz, 10 μs

  • delay –> frequency-to-phase conversion
  • works at any frequency
  • long delay (microseconds) is necessary for high sensitivity
  • the delay line must be an optical fiber

fiber: attenuation 0.2 dB/km, thermal coeff. 6.8 10-6/K cable: attenuation 0.8 dB/m, thermal coeff. ~ 10-3/K

Rubiola, Salik, Huang, Yu, Maleki, JOSA-B 22(5) p.987–997 (2005)

Φ(s) = Hϕ(s)Φi(s)

Laplace transforms

Sy(f) = |Hy(f)|2 Sϕ i(s)

|Hϕ(f)|2 = 4 sin2(πfτ) |Hy(f)|2 = 4ν2 f 2 sin2(πfτ) 10 GHz, 10 μs

Σ

kϕ −s

e

τ Φo(s) Φi(s) V

  • (s) kϕΦo(s)

= Φo(s)

τ −s

(1−e )Φi(s) = mixer

+

detector

mW 10

Pλ τd = 1.. 100 µ s

EOM

90° adjust τ∼ _0 laser µm 1.55

mW 100

_ ∼ τd 0 20−40 dB R0 52 dB FFT analyz. (t) vo

  • ut

(0.2−20 km) power ampli input microwave (calib.) phase

The short arm can be a microwave cable or a photonic channel Laplace transforms

Qeq = πν0τ

slide-21
SLIDE 21

21

  • The instrument noise scales as 1/τ, yet the

blue and black plots overlap magenta, red, green => instrument noise blue, black => noise of the sapphire

  • scillator under test
  • We can measure the 1/f3 phase noise

(frequency flicker) of a 10 GHz sapphire

  • scillator (the lowest-noise microwave
  • scillator)
  • Low AM noise of the oscillator under test is

necessary

Att FFT

DC JDS Uniphase JDS Uniphase 1,5 µm = Contrôleur de polarisation Photodiode DSC40S Déphaseur Ampli DC Analyseur FFT (HP 3561A)

Coupleur 10 dB

Ampli RF

3dB

Ampli AML 8-12GHz

LO RF 5 dBm 10 dBm

ISO ISO

Fibre 2 Km

laser EOM SiGe ampli phase 2 km sapphire oscillator

Measurement of a sapphire oscillator

Volyanskiy & al., JOSAB (in press). Also arXiv:0807.3494v1 [physics.optics] July 2008.

slide-22
SLIDE 22

Dual-channel (correlation) measurement

22

Improvements

Derives from: E. Salik, N. Yu, L. Maleki, E. Rubiola, Proc. Ultrasonics-FCS Joint Conf., Montreal, Aug 2004 p.303-306

  • Understanding flicker (photodetectors and amplifiers)
  • SiGe technology provides lower 1/f phase noise
  • CATV laser diodes exhibit lower AM/FM noise
  • Low Vπ EOMs show higher stability because of the lower RF power
  • Optical fiber sub-mK temperature controlled

Volyanskiy & al., JOSAB (in press) and arXiv:0807.3494v1 [physics.optics] July 2008.

!'" !(" !%" !#" !"" '" (" %" #" ;<9/0=.*17.1>*-?@17.1A=9B.19C.011-/1*.@9*717.1!"DB12E?>*.1#FG5 6A.01H.*IE<.J1K!"F3419C.01-/1*.@9*717.1#"DB12E?>*.1%FG51 J.Cussey 20/02/07 Mesure200avg.txt

–20 –180 –40 –60 –80 –160 –140 –120 –100 101 102 103 104 105

Fourier frequency, Hz S(f), dBrad2/Hz residual phase noise (cross-spectrum), short delay (0), m=200 averaged spectra, unapplying the delay eq. with =10 s (2 km)

J.Cussey, feb 2007

y = 10–12 baseline

FFT average effect FFT average effect FFT average effect

slide-23
SLIDE 23

Delay-line oscillator – operation

23

Σ

+ + A

free noise

  • V’(s)

Vo(s) V

i(s)

initial conditions, noise, or locking signal βf(s) selector βd

τd

e−s = (s) delay

τd

e−s model output

  • utput
  • scillator

true in practice, delay + selector delay = (s) β

+2π/τd . . . . H(s) . . . . σ l=+3 l=+2 l=+1 l=0 l=−1 l=−2 l=−3 . . . . . . . . j ω τd ln(A) 1 +6π/τd −6π/τd +4π/τd −2π/τd −4π/τd

1 2 3 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20

f * tau transfer function |H(jf)|^2 A=1 A=0.75 A=0.71 A=0.65 A=0.50 A=0.30

file le−calc−hdly−flt src allplots−leeson

delay−line loop, no selection filter

General feedback theory H (s) = Vo(s) Vi(s) = 1 1 A (s) Delay-line oscillator H (s) = 1 1 A e

sτd

Location of the roots s

l = 1

  • d

ln(A ) + j2

  • d

l integer l( , )

  • E. Rubiola, Phase Noise and Frequency Stability in Oscillators, Cambridge 2008, ISBN 978-0521-88677-2

Barkhausen condition for oscillation: Aß = 1

slide-24
SLIDE 24

Delay-line oscillator – phase noise

24

1000 10000 1e+05 1e+06 0.01 0.1 1 10 100 1000 10000 1e+05 1e+06 1e+07 1e+08

phase noise |H(jf)|^2 delay−line oscillator with selector

frequency, Hz

parameters: tau = 2E−5 s m = 2E5 nu_m = 10 GHz Q = 2000

file le−calc−dly−hphase src allplots−leeson

1

Σ

+ +

(s) Ψ (s) Φ free noise (s) B

f

τ 1 / (1+s ) = (s) B

τd

e−s = delay (s) B

τd

e−s = delay phase noise input phase noise

  • utput

selector in practice, delay + selector ω m σ µ=0 µ=−1 µ=−2 . . . . . . . . µ=+2 µ=+1 = j ωm 2Q µ m τd Δω = − 2Q µ m τd Δω = − 2Q ωm (2π/τd) 2Q2 µ m

2

τd σ = − (2π/τd) µ (2π/τd) µ

General feedback theory H(s) = Φ(s) Ψ(s) = 1 1 − B(s) Delay-line oscillator H(s) = 1 + sτf 1 + sτf − e−sτd Location of the roots sµ = −2Q2 τd µ m 2 + j 2π τd µ − 2Q τd µ m

  • E. Rubiola, Phase Noise and Frequency Stability in Oscillators, Cambridge 2008, ISBN 978-0521-88677-2
slide-25
SLIDE 25

Delay-line oscillator

25 fL = 1 4π2τ 2 fL = ν0 2Q Qeq = πν0τ Qeq=3×105 ← L=4km Sϕ(f) ≃ f 2

L

f 2 Sψ(f) for f ≪ fL fL=8kHz

Leeson formula

σy ≃ 2.9×10−12

10–11 Allan deviation h−1 = b−3/ν2 6.3×10–24 8.8×10–24 σ2

y = 2 ln(2) h−1

b–3 = 6.3×10–4 (–32 dB)

slide-26
SLIDE 26

Delay-line oscillator - measured noise

26 expected phase noise b–3 ≈ 6.3×10–4 (–32 dB)

  • 1.310 nm DFB CATV laser
  • Photodetector DSC 402 (R = 371 V/W)‏
  • RF filter ν0 = 10 GHz, Q = 125
  • RF amplifier AML812PNB1901 (gain +22dB)‏
  • ur OEO

b–3=10–3 (–30dB)

Agilent E8257c, 10 GHz, low-noise opt. Wenzel 501-04623 OCXO 100 MHz

  • mult. to 10 GHz

101 102 103 104 105 –20 –40 –60 –80 –160 –140 –120 –100

S(f), dBrad2/Hz

Phase noise of the opto-electronic oscillator (4 km)

frequency, Hz

  • E. Rubiola, apr 2008

OEO: Kirill Volyanskiy, may 2007

slide-27
SLIDE 27

Optical-fiber 10 GHz oscillator

!"# $%&' ()%&' *+,&-. ,%/+. 0.12 3425.,&'+6 78.96%:%/;. 5<6= (&6%&),+. <5=%>&,.><45,+6 3?<=<9+=+>=<6 78.&25,%@%+6 78.8%,=+6 3<,&6%A&=%</. ></=6<,,+6 *+,&-. ,%/+. B.12 3?<=<9+=+>=<6

C C

D==

Kiryll Volyanskiy, jan 2008

27

  • use positive feedback with a short cable (3-5 ns) in the feedback path to implement

the mode selector filter

  • the positive feedback also increase the amplifier gain

(AML SiGe parallel amplifiers exhibits lowest flicker, but low have gain 22 dB)

  • use the 2-km (10 µs) path to eliminate the 50-kHz noise peak due to the 4-km (20 µs)
  • the microwave power is changed by adjusting the laser power
  • high noise figure, due to the two power splitters/combiners
slide-28
SLIDE 28

The oscillator phase noise minima are 6 dB lower than b0=N/P0 (white noise) m = 0.725 (Prf=11 dBm) (Sφ)min = –142 dB F = 10 dB (incl. couplers) η = 0.6 νl = 194 THz (Sϕ)min = 8 m2

  • FkBT0

R0 hνl qη 2 1 P

2 l

+ 2 hνl η 1 P l

  • 10

1

10

2

10

3

10

4

10

5

160 140 120 100 80 60 40 20 Frequency (Hz) S (dB rad2/Hz) 8dBm 9dBm 11dBm

b–2 = –50.5 dBrad2/Hz (8 dBm) b–3 = –26 dBrad2/Hz b–4 = –7 dBrad2/Hz b–2 = –53 dBrad2/Hz (9 dBm) b–2 = –57 dBrad2/Hz (11 dBm) file 923-kirill-oeo E.R & K.Voliansky, jan 2008 (S)min = –142 dBrad2/Hz (11 dBm) the power is measured at the amplifier output

we get P0 = 6.4 μW (-22 dBm) Pl ≈ 0.71 mW There is room for engineering

Regenerative optical-fiber 10 GHz oscillator

Prf is given, thus V0 =(2RP)1/2 Vπ is estimated (4.5 V at 10 GHz) Use m = 2J1 πV0 Vπ

  • P, dBm

Vp, V πV0/Vπ m 11 1.122 0.860 0.783 9 0.891 0.683 0.644 8 0.794 0.6.09 0.581

Feeding the available data in the model Get 28

slide-29
SLIDE 29

3 – Electronic and optical components

slide-30
SLIDE 30

Flicker in electronic & optical components

30

stopband

  • utput bandwidth

stopband

  • utput bandwidth

near-dc flicker

no carrier

S(f) f S(f) f

noise up-conversion

near- dc noise

expand and select the ω0 terms carrier

vi(t) = Vi ejω0t + n′(t) + jn′′(t)

non-linear (parametric) amplifier

vo(t) = Vi

  • a1 + 2a2
  • n′(t) + jn′′(t)
  • ejω0t

get AM and PM noise

α(t) = 2 a2 a1 n′(t) ϕ(t) = 2 a2 a1 n′′(t)

The AM and the PM noise are independent of Vi , thus of power

vo(t) = a1vi(t) + a2v2

i (t) + . . . substitute

(careful, this hides the down-conversion)

the parametric nature of 1/f noise is hidden in n’ and n”

ω0 = ?

no flicker

ω0

The noise sidebands are proportional to the input carrier

near-dc noise

There is also a linear parametric model, which yields the same results

slide-31
SLIDE 31

31

=−10dBm Pin=−5dBm Pin=0dBm −165 −160 −155 −150 −145 −140 −135 −130 −125 −120 f, Hz Phase noise, dBrad 2/Hz SiGe LPNT32

bias 2V, 20 mA

=−15dBm

  • R. Boudot 2006

100 1 10 1000 10000 100000 −175 P −170

in

Pin

Amplifier flicker noise – experiments

  • S (f), dB.rad /Hz

2

23 Jan 07 Avantek UTC573, 10 MHz 56 dB

NMS floor

f, Hz

120 130 140 150 160 170 180 1 10 100 1000 10000 100000

1Amp Pin=5dBm 3Amps Pin= 24dBm 2Amps Pin= 5dBm

  • The 1/f phase noise b–1 is about independent of

power

  • The white noise b0 scales up/down as 1/P0, i.e., the

inverse of the carrier power

  • Describing the 1/f noise in terms of fc is misleading

because fc depends on the input power

  • The expected flicker of a cascade increases by:

3 dB, with 2 amplifiers 5 dB, with 3 amplifiers

S (f), dB.rad /Hz

2

  • 2 parallel amplifiers

Pin= −5 dBm Single amplifier Pin= −6 dBm JS2, 10 GHz 15 Feb 2007

f, Hz

50 70 90 110 130 150 170 1 10 100 1000 10000 100000

Phase noise vs. power Phase noise of cascaded amplifiers Phase noise of paralleled amplifiers

  • Connecting two amplifiers in parallel, the phase-

noise flicker is expected to decrease by 3 dB

Regenerative amplifiers

  • Phase noise increase as the squared gain because

the noise source at each roundtrip is correlated

slide-32
SLIDE 32

32

Photodetector 1/f noise

r(t) iso iso P

90° ° 0° 90°

RF LO IF

FFT analyz. power meter

=6dB (detection of or )

  • phase

PLL

synth. 9.9GHz MHz 100 power ampli laser YAG 1.32 µ m

EOM

(3dBm)

s(t)

  • (26dBm)

hybrid

g=37dB g’=52dB (carrier suppression)

phase & aten.

50% coupler

infrared v(t) 22dBm

(13dBm)

monitor

  • utput

photodiodes under test microwave neardc

The noise of the ∑ amplifier is not detected [Rubiola, Salik, Yu, Maleki, Electron. Lett. 39(19) p.1389-1390 (2003) ]

photodiode Sα(1 Hz) Sϕ(1 Hz) estimate uncertainty estimate uncertainty HSD30 −122.7

−7.1 +3.4

−127.6

−8.6 +3.6

DSC30-1K −119.8

−3.1 +2.4

−120.8

−1.8 +1.7

QDMH3 −114.3

−1.5 +1.4

−120.2

−1.7 +1.6

unit dB/Hz dB dBrad2/Hz dB Rubiola, Salik, Yu, Maleki, MTT 54(2) p.816-820, Feb 2006

slide-33
SLIDE 33

33

Photodetector 1/f noise

  • the photodetectors we measured are similar in

AM and PM 1/f noise

  • the 1/f noise is about -120 dB[rad2]/Hz
  • ther effects are easily mistaken for the

photodetector 1/f noise

  • environment and packaging deserve attention

in order to take the full benefit from the low noise of the junction

W: waving a hand 0.2 m/s, 3 m far from the system B: background noise P: photodiode noise S: single spectrum, with optical connectors and no isolators B: background noise P: photodiode noise A: average spectrum, with optical connectors and no isolators B: background noise P: photodiode noise F: after bending a fiber, 1/f noise can increase unpredictably B: background noise P: photodiode noise

Rubiola, Salik, Yu, Maleki, MTT 54(2) p.816-820, Feb 2006

slide-34
SLIDE 34

Microwave optical link

34

f

Sϕ(f)

10–32 @ 1 Hz

τ

4 x 1 0–17 /

  • σy(τ)

f

10–12 rad2/Hz @ 1 Hz

Sy(f)

Sy(f) = f 2 ν2 Sϕ(f) σ2

y(τ) = [1.038 + 3 ln(2πfHτ)]

h1 (2π)2 1 τ 2 b−1/f h1f

  • Let’s be optimistic: a 10 GHz link is

limited by the 1/f phase noise of a single component, –120 dBrad2/Hz @ f=1 Hz

  • Well known rules give σy(τ) = 4x10–17/τ
  • Realistically, –100 dBrad2/Hz @ f=1 Hz

yields σy(τ) = 4x10–16/τ

slide-35
SLIDE 35

4 – AM noise and RIN

slide-36
SLIDE 36

Amplitude noise & laser RIN

36

  • E. Rubiola, the measurement of AM noise, dec 2005

arXiv:physics/0512082v1 [physics.ins-det]

monitor source under test dual channel FFT analyzer vb va Pb Pa power meter monitor R0 R0 Pa Pb

coupler

power meter

coupler

source under test R R va vb dual channel FFT analyzer power meter microwave

  • ptical

monitor dc

dc power meter

coupler coupler

source under test Pb Pa R R dual channel FFT analyzer va vb monitor

  • ptical

−123.1 10 102 103 104 105

Fourier frequency, Hz

avg 2100 spectra = −10.2 dBm P

Wenzel 501−04623E 100 MHz OCXO

(f ) Sα

dB/Hz −163.1 −153.1 −143.1 −133.1

  • In PM noise measurements, one can validate the instrument by

feeding the same signal into the phase detector

  • In AM noise this is not possible without a lower-noise reference
  • Provided the crosstalk was measured otherwise, correlation

enables to validate the instrument

AM noise of RF/microwave sources Laser RIN AM noise of photonic RF/microwave sources

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Kirill Volyanskiy

slide-37
SLIDE 37

AM noise of some sources

37 source h−1 (flicker) (σα)floor Anritsu MG3690A synthesizer (10 GHz) 2.5×10−11 −106.0 dB 5.9×10−6 Marconi synthesizer (5 GHz) 1.1×10−12 −119.6 dB 1.2×10−6 Macom PLX 32-18 0.1 → 9.9 GHz multipl. 1.0×10−12 −120.0 dB 1.2×10−6 Omega DRV9R192-105F 9.2 GHz DRO 8.1×10−11 −100.9 dB 1.1×10−5 Narda DBP-0812N733 amplifier (9.9 GHz) 2.9×10−11 −105.4 dB 6.3×10−6 HP 8662A no. 1 synthesizer (100 MHz) 6.8×10−13 −121.7 dB 9.7×10−7 HP 8662A no. 2 synthesizer (100 MHz) 1.3×10−12 −118.8 dB 1.4×10−6 Fluke 6160B synthesizer 1.5×10−12 −118.3 dB 1.5×10−6 Racal Dana 9087B synthesizer (100 MHz) 8.4×10−12 −110.8 dB 3.4×10−6 Wenzel 500-02789D 100 MHz OCXO 4.7×10−12 −113.3 dB 2.6×10−6 Wenzel 501-04623E no. 1 100 MHz OCXO 2.0×10−13 −127.1 dB 5.2×10−7 Wenzel 501-04623E no. 2 100 MHz OCXO 1.5×10−13 −128.2 dB 4.6×10−7 worst best